The present invention relates to a switching power circuit adapted for use as a power supply in various electronic apparatus.
There are widely known switching power circuits of a type employing a switching converter such as a flyback converter or a forward converter. Since such a switching converter performs its switching operation with rectangular waves, there exists a limit in suppression of switching noise. And it is also obvious that, due to the operating characteristic thereof, some restriction is unavoidable in improving the power conversion efficiency.
In view of the points mentioned above, a variety of switching power circuits employing various resonant converters have already been proposed by the present applicant. A resonant converter is capable of attaining a high power conversion efficiency with facility and realizing low noise as the switching operation is performed with sinusoidal waves. And it is further possible to achieve another merit that the circuit can be constituted of a relatively small number of component parts.
FIG. 6 is a circuit diagram showing a conventional switching power circuit of a configuration based on the known invention filed previously by the present applicant.
In the power circuit shown in this diagram, a full-wave rectifier circuit consisting of a bridge rectifier Di and a smoothing capacitor Ci is provided as a rectifier smoothing circuit for obtaining a DC input voltage from a commercial alternating power supply (alternating input voltage VAC), wherein a rectified smoothed voltage Ei corresponding to one-fold level of the alternating input voltage VAC is generated.
As a switching converter for intermittently turning on and off the input rectified smoothed voltage Ei (DC input voltage), there is provided a voltage resonant converter which comprises a switching element Q1 of one transistor to perform its switching operation in a single end form.
The voltage resonant converter employed here adopts a separately excited structure, and the switching element Q1 consists of a MOS-FET for example. The drain of this switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci via a primary winding N1 of the insulating converter transformer PIT, and its source is connected to a primary-side ground.
A parallel resonance capacitor Cr is connected between the drain and source of the switching element Q1. The capacitance of this parallel resonance capacitor Cr and a leakage inductance obtained in the primary winding N1 of the insulating converter transformer PIT constitute a primary parallel resonance circuit. And a resonance action is caused by the parallel resonance circuit in accordance with the switching operation of the switching element Q1, so that such switching operation of the switching element Q1 becomes a voltage resonance type.
Further a clamp diode DD consisting of a body diode is connected in parallel between the drain and source of the switching element Q1, thereby forming a path of a clamp current which flows during the off-time of the switching element.
In this case, the drain of the switching element Q1 is connected to an oscillation circuit 41 in a switching driver 10B which will be described next. The drain output supplied to the oscillation circuit 41 is used for variably controlling the switching on-time in control of the switching frequency as will be mentioned later.
The switching element Q1 is driven by the switching driver 10B which is integrally equipped with the oscillation circuit 41 and a drive circuit 42, and the switching frequency is variably controlled for execution of constant voltage control. The switching driver 10B in this case is provided as a single integrated circuit (IC) for example.
The switching driver 10B is connected to the line of a rectified smoothed voltage Ei via a start resistor Rs, and at a power supply start time for example, a source voltage is impressed via the start resistor Rs to thereby start the switching driver 10B.
The oscillation circuit 41 in the switching driver 10B performs oscillation to thereby generate an oscillation signal and then outputs the same. Subsequently in the drive circuit 42, this oscillation signal is converted into a driving voltage and then is outputted to the gate of the switching element Q1. Thus, the switching element Q1 performs its switching operation based on the oscillation signal generated in the oscillation circuit 41. Therefore, the switching frequency of the switching element Q1 and the on/off duty ratio in one switching period are determined depending on the oscillation signal generated in the oscillation circuit 41.
The oscillation circuit 41 performs its operation of changing the oscillation signal frequency (switching frequency fs) on the basis of the level of a secondary DC output voltage EO which is supplied via a photo coupler 30 as will be mentioned later. And simultaneously with the operation of changing the switching frequency fs, the oscillation circuit 41 further serves to control the oscillation signal waveform in such a manner that the on-time TON (conduction angle) of the switching element Q1 is changed while the off-time TOFF of the switching element Q1 is maintained constant. Consequently, the secondary DC output voltage EO can be stabilized due to such operation of the oscillation circuit 41, as will be described later.
The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As shown in FIG. 8, the insulating converter transformer PIT has an EE-shaped core where E-shaped cores CR1 and CR2 composed of ferrite for example are combined with each other in such a manner that magnetic legs thereof are opposed mutually, and the primary winding N1 and the secondary winding N2 thereof are coiled in a split state respectively by the use of a split bobbin B with regard to the center magnetic leg of the EE-shaped core. And a gap G is formed to the center magnetic leg as shown in the diagram, whereby coarse coupling is attained with a required coupling coefficient.
The gap G can be formed by shaping the center magnetic leg of each of the E-shaped cores CR1 and CR2 to be shorter than the two outer magnetic legs thereof. The coupling coefficient k is set as, e.g., k.apprxeq.0.85 suited to attain coarse coupling, hence avoiding a saturated state correspondingly thereto.
As shown in FIG. 6, the end of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching element Q1, while the beginning of the primary winding N1 is connected to the positive terminal (rectified smoothed voltage Ei) of the smoothing capacitor Ci. Therefore, when the switching output of the switching element Q1 is supplied to the primary winding N1, there is generated an alternating voltage of a period corresponding to the switching frequency.
On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that a parallel resonance circuit is formed by a combination of the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. And the alternating voltage induced in the secondary winding N2 by this parallel resonance circuit is a resonance voltage, i.e., a voltage resonance action is caused on the secondary side.
More specifically, this power circuit has, on its primary side, a parallel resonance circuit for turning the the switching operation into a voltage resonance type, and also has, on its secondary side, another parallel resonance circuit for producing a voltage resonance action. In this specification, the switching converter of a configuration equipped with resonance circuits on its primary and secondary sides as mentioned above will be referred to as "composite resonant switching converter".
On the secondary side of the power circuit formed as described above, a rectifier smoothing circuit consisting of a bridge rectifier DBR and a smoothing capacitor CO is provided to obtain a secondary DC output voltage EO. That is, in this configuration, full-wave rectification is performed by the bridge rectifier DBR on the secondary side. In this case, as a resonance voltage is supplied from the secondary parallel resonance circuit, the bridge rectifier DBR generates a secondary DC output voltage EO which is substantially equal in level to the alternating voltage induced in the secondary winding N2.
In a state where the primary side and the secondary side are isolated from each other via a photo coupler 30 with respect to direct current, the secondary DC output voltage EO is inputted also to the oscillation circuit 41 in the primary switching driver 10B.
As for the secondary-side operation in the insulating converter transformer PIT, the mutual inductance M regarding the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes either +M or -M depending on the relation of the polarities (winding directions) of the primary winding N1 and the secondary winding N2 to the connection of the rectifying diodes DO (DO1, DO2), and also depending on the polarity change of the alternating voltage induced in the secondary winding N2.
For example, if the circuit is equivalent to one shown in FIG. 9A, the mutual inductance becomes +M. Meanwhile, if the circuit is equivalent to another shown in FIG. 9B, the mutual inductance becomes -M.
Applying the above to the secondary-side operation shown in FIG. 6, when the alternating voltage obtained in the secondary winding N2 is positive for example, it is supposed that the operation with the rectified current flowing in the bridge rectifier circuit DBR is performed in the +M (forward) mode. Meanwhile, when the alternating voltage obtained in the secondary winding N2 is negative contrary to the above, it is supposed that the operation with the rectified current flowing in the bridge rectifier diode DBR is performed in the -M (flyback) mode. Thus, every time the alternating voltage obtained in the secondary winding N2 is turned to be positive or negative, the operation mode is changed to +M or -M relative to the mutual inductance.
In this configuration, the power increased by the action of the primary parallel resonance circuit and the secondary parallel resonance circuit is supplied to the load, so that the power to be supplied to the load is also increased correspondingly thereto to consequently enhance the increase rate of the maximum load power.
Such correspondence to the load condition can be realized due to the improved situation where a saturated state is not reached readily because of the coarse coupling attained by a required coupling coefficient with the gap G formed in the insulating converter transformer PIT, as explained previously with reference to FIG. 8. For example, in case the gap G is not existent in the insulating converter transformer PIT, the operation will be abnormal with a high probability as the insulating converter transformer PIT is placed in its saturated state during the flyback, whereby proper execution of the aforementioned full-wave rectification is rendered considerably difficult.
The stabilizing operation in the circuit of FIG. 6 is performed in the following manner.
As mentioned, the secondary DC output voltage EO is supplied via the photo coupler 30 to the oscillation circuit 41 in the primary switching driver 10B. Subsequently in the oscillation circuit 41, the oscillation signal frequency is changed in accordance with the level change of the secondary DC output voltage EO thus supplied, and then the oscillation signal is outputted. Due to this operation that changes the switching frequency of the switching element Q1, the resonance impedances of the primary voltage resonant converter and the insulating converter transformer PIT are also changed to consequently change the energy transmitted to the secondary side of the insulating converter transformer PIT. As a result, the secondary DC output voltage EO is maintained constant under control at a required level, hence stabilizing the power supply.
In the power circuit shown in FIG. 6, as described already, the switching frequency is changed in the oscillation circuit 41 by variably controlling the on-time TON of the switching element Q1 while keeping the off-time TOFF thereof fixed. More specifically, in this power circuit, its constant voltage control action is executed in a manner to variably control the switching frequency to thereby achieve control of the resonance impedance to the switching output, and simultaneously another control action is executed with regard to the conduction angle control (PWM control) of the switching element in the switching period. Such composite control action is realized by a set of control circuitry. In this specification, such composite control is termed "composite control system".
FIG. 7 shows another conventional power circuit of a configuration based on the contents proposed previously by the present applicant. In this diagram, any component parts corresponding to those in FIG. 6 are denoted by the same reference numerals or symbols, and a repeated explanation thereof is omitted here.
On the primary side of the power circuit shown in FIG. 7, a self-excited structure is provided as a voltage resonant converter circuit where a single end operation is performed by a switching element Q1 of one transistor. In this case, a high withstand-voltage bipolar transistor (BJT: junction transistor) is employed as the switching element Q1.
The base of the switching element Q1 is connected to the positive side of a smoothing capacitor Ci (rectified smoothed voltage Ei) via a base current limiting resistor RB and a starting resistor RS, so that a base current at the start is obtained from a rectifier smoothing line. And a series resonance circuit for self-excited oscillation driving, which consists of a series connection circuit of a driving coil NB, a resonance capacitor CB and a base current limiting resistor RB, is connected between the base of the switching element Q1 and a primary-side ground.
A path of a clamp current flowing during the off-time of the switching element Q1 is formed by a clamp diode DD inserted between the base of the switching element Q1 and the negative terminal (primary-side ground) of the smoothing capacitor Ci. Meanwhile, the collector of the switching element Q1 is connected to one end of the primary winding N1 of an insulating converter transformer PIT, and the emitter thereof is grounded.
A parallel resonance capacitor Cr is connected in parallel between the collector and emitter of the switching element Q1. In this case also, the capacitance of the parallel resonance capacitor Cr itself and a leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT constitute a primary parallel resonance circuit of the voltage resonant converter.
An orthogonal control transformer PRT shown in this diagram is a saturable reactor where a resonance current detection coil ND, a driving coil NB and a control coil NC are wound. This orthogonal transformer PRT is provided for driving the switching element Q1 and also for executing constant voltage control.
In the structure of this orthogonal control transformer PRT, although not illustrated, two double U-shaped cores having four magnetic legs form a solid core where the ends of the respective magnetic legs are mutually joined. And a resonance current detection coil ND and a driving coil NB are wound around two predetermined magnetic legs of the solid core in the same direction, and further a control coil NC is wound orthogonally to the resonance current detection coil ND and the driving coil NB.
In this case, the resonance current detection coil ND of the orthogonal control transformer PRT is inserted in series between the positive terminal of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching output of the switching element Q1 is transmitted to the resonance current detection coil ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection coil ND is induced in the driving coil NB through transformer coupling, hence generating an alternating voltage as a driving voltage in the driving coil NB. This driving voltage is delivered as a driving current from the series resonance circuit (NB, CB), which constitutes a self-excited oscillation driving circuit, to the base of the switching element Q1 via the base current limiting resistor RB. Consequently, the switching element Q1 performs its switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.
The insulating converter transformer PIT included in the circuit of FIG. 7 is structurally the same as the one described previously with reference to FIG. 8, so that the primary side and the secondary side thereof are in a state of coarse coupling.
Also on the secondary side of this insulating converter transformer PIT in the circuit of FIG. 7, a secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2 to thereby constitute a secondary parallel resonance circuit, so that the configuration of a composite resonant switching converter is attained in this power circuit as well.
On the secondary side of this power circuit, a half-wave rectifier circuit consisting of a single diode DO and a smoothing capacitor CO is provided to the secondary winding N2, wherein a secondary DC output voltage EO is obtained by half-wave rectification in its forward operation alone. In this case, the secondary DC output voltage EO is branched and inputted also to a control circuit 1, wherein the DC output voltage EO is used as a detection voltage.
In the control circuit 1, the level of a control current (direct current) flowing in a control coil NC is changed in accordance with a change of the secondary DC output voltage level EO, thereby variably controlling the inductance LB of the driving coil NB wound around the orthogonal control transformer PRT. Consequently, the resonance condition of the series resonance circuit is changed in a self-excited oscillation driving circuit formed inclusively of the inductance LB of the driving coil NB for the switching element Q1. The above is an operation for changing the switching frequency of the switching element Q1 to thereby stabilize the secondary DC output voltage. In such constant voltage control configuration equipped with the orthogonal control transformer PRT, the primary switching converter is formed into a voltage resonant type, whereby there is achieved a composite control action which executes variable control of the switching frequency and, simultaneously therewith, conduction angle control (PWM control) of the switching element in the switching period.
FIGS. 10A through 10F are waveform charts showing the operation of the primary voltage resonant converter in the power circuits of FIGS. 6 and 7. FIGS. 10A through 10C represent the operation performed under the conditions of AC input voltage VAC=100V and maximum load power Pomax =200 W; and FIGS. 10D through 10F represent the operation performed under the conditions of AC input voltage VAC=100V and minimum load power Pomin=0 W.
In response to a switching operation of the switching element Q1, a resonance action of the primary parallel resonance circuit is produced during the off-time TOFF of the switching element Q1. Consequently, a parallel resonance voltage V1 obtained across the parallel resonance capacitor Cr is such as shown in FIGS. 10A and 10D where a sine-wave resonance pulse is generated during the time TOFF.
Since such a parallel resonance action is produced during the time TOFF, a parallel resonance current Icr flowing in the parallel resonance capacitor Cr becomes such as shown in FIGS. 10C and 10F where the current of a substantially sinusoidal wave flows with a transition from a positive direction to a negative direction during the time TOFF.
As obvious from comparison of FIG. 10A with FIG. 10D, the switching frequency fs is so controlled as to become higher with a decrease of the load power Po, and the switching frequency fs (switching period) is varied by changing the on-time TON of the switching element Q1 while keeping the off-time TOFF thereof fixed. That is, the operation conforming with the aforementioned composite control system is represented in the diagrams.
In the configuration of the voltage resonant converter shown in FIGS. 6 and 7, the level of the parallel resonance voltage V1 changes in accordance with a variation of the load power. For example, it becomes 550 Vp when the maximum load power Pomax=200 W, or 300 Vp when the minimum load power Pomin=0 W. That is, the parallel resonance voltage V1 tends to rise with an increase of the load power.
As shown in FIGS. 10B and 10E, the switching output current IQ1 flowing in the drain or collector of the switching element Q1 flows with a zero level during the time TOFF or flows with the shown waveform during the time TON. The level of this switching output current IQ1 also tends to rise with an increase of the load power Po. According to these diagrams for example, it becomes 3.8 A when the maximum load power Pomax=200 W, or 1A when the minimum load power Pomin=0 W.
Regarding the characteristics of the power circuits shown in FIGS. 6 and 7, FIG. 11 represents variation characteristics of the switching frequency fs, the times TOFF and TON within one switching period, and the parallel resonance voltage V1 to the AC input voltage VAC when the maximum load power Pomax=200 W.
As shown graphically in FIG. 11, first the switching frequency fs is changed approximately in a range of fs=110 kHz to 140 kHz with respect to a variation range of the AC input voltage VAC=90V to 140V. This graph indicates that any variation of the secondary DC output voltage EO is stabilized in accordance with a variation of the DC input voltage. Upon variation of the AC input voltage VAC, the switching frequency is raised under control in accordance with a rise of the AC input voltage VAC.
As for the times TOFF and TON in one switching period, the time TOFF is kept fixed regardless of the switching frequency fs, while the time TON is lowered on a quadratic curve in accordance with a rise of the switching frequency fs. Thus, the graph also indicates that the switching frequency control is executed in conformity with the composite control system.
Meanwhile the parallel resonance voltage V1 is changed in accordance with a variation of the commercial AC power VAC and, as shown graphically, its level is varied to be higher with a rise of the AC input voltage VAC.
As shown in FIGS. 6 and 7 for example, in the power circuit so constituted as to stabilize the secondary DC output voltage by the composite control system, the peak level of the parallel resonance voltage V1 changes in accordance with load conditions and variations of the AC input voltage VAC, as shown in FIGS. 10A, 10B and 11 also. Particularly when the level of the AC input voltage VAC obtained from, e.g., a 100V commercial AC power supply has reached 140V in a heavy load state close to the maximum load power, then the parallel resonance voltage V1 rises maximally to 700 Vp, as shown in FIG. 11.
Therefore, with regard to the parallel resonance capacitor Cr and the switching element Q1 to which the parallel resonance voltage V1 is applied, these component parts need to meet a withstand voltage requisite of 800V when used with a 100V commercial AC power supply, or to meet a withstand voltage requisite of 1200V when used with a 200V commercial AC power supply. Consequently, it is unavoidable that both the parallel resonance capacitor Cr and the switching element Q1 become larger in size, and the cost is rendered higher as well.
The switching element has such a feature that its characteristic is deteriorated with a structural alteration to attain a higher voltage withstand performance. For this reason, if the switching element Q1 is so selected as to meet a higher withstand voltage requisite, its power loss resulting from the switching operation is increased to consequently bring about deterioration of the power conversion efficiency.
In the case of adopting a configuration to stabilize the secondary DC output voltage by the composite control system, if there occurs a fault that the secondary load is shorted, the controller functions to lower the switching frequency. In a state where the switching frequency becomes lower, as obvious from the waveform charts of FIGS. 10A through 10F, the on-time TON of the switching element is rendered longer to eventually raise the voltage (V1) applied to the switching element Q1 and the parallel resonance capacitor Cr and also to increase the currents (IQ1, Icr) flowing therein.
Accordingly, it becomes necessary to prepare, as a countermeasure against occurrence of a load shorted fault, an overcurrent protection circuit and an overvoltage protection circuit for protecting the switching element by limiting any high-level voltage and current generated in such a fault. And the existence of such overcurrent and overvoltage protection circuits further impedes improvement in down-sizing and cost reduction of the entire circuits.